Linear phase FIR sinc filter with multiplexing

ABSTRACT

A phase shifter is implemented using a polyphase filter. The filter is preferably a linear phase Finite Impulse Response (FIR) filter. The amount of delay imparted by the phase shifter is determined by a particular set of coefficients selected from a plurality of such coefficients. Storage requirements are reduced by taking advantage of symmetries in the coefficients for the filters. Memory requirements are further reduced by partitioning the polyphase filter into two polyphase filters and using one to set a rough delay amount and the other to set a fine delay amount between rough delay amount settings. The particular amount of delay may be set by an external synchronization signal.

CROSS-REFERENCES TO RELATED APPLICATIONS

The invention disclosed herein is related to application Ser. No.09/153,863, filed Sep. 16, 1998, by inventors Joel Page, Edwin De Angel,Wai Laing Lee, Lei Wang, Hong Zheng and Chung-Kai Chow and entitled “APOLYPHASE FILTER WITH SELECTIVE PHASE SHIFTING.”

The invention disclosed herein is related to application Ser. No.09/153,862, filed Sep. 16, 1998, by inventors Joel Page, Edwin De Angel,Wai Laing Lee, Lei Wang, Hong Zheng and Chung-Kai Chow and entitled “ASINC FILTER WITH SELECTIVE DECIMATION RATIOS.”

The invention disclosed herein is related to application Ser. No.09/153,860, filed Sep. 16, 1998, by inventors Joel Page, Edwin De Angel,Wai Laing Lee, Lei Wang, Hong Zheng and Chung-Kai Chow and entitled “ASINC FILTER USING TWISTING SYMMETRY.”

The invention disclosed herein is related to application Ser. No.09/154,242, filed Sep. 16, 1998, by inventors Joel Page, Edwin De Angel,Wai Laing Lee, Lei Wang, Hong Zheng, Chung-Kai Chow and entitled“NETWORK SYNCHRONIZATION.”

The invention disclosed herein is related to application Ser. No.09/153,861, filed Sep. 16, 1998, by inventors Joel Page, Edwin De Angel,Wai Laing Lee, Lei Wang, Hong Zheng and Chung-Kai Chow and entitled“CLOCK ALIGNMENT FOR REDUCED NOISE AND EASY INTERFACING.”

The invention disclosed herein is related to application Ser. No.09/153,869, filed Sep. 16, 1998, by inventors Joel Page, Edwin De Angel,Wai Laing Lee, Lei Wang, Hong Zheng and Chung-Kai Chow and entitled “ACHIP ARCHITECTURE FOR DATA ACQUISITION.”

The invention disclosed herein is related to application Ser. No.09/153,867, filed Sep. 16,1998, by inventors Joel Page, Edwin De Angel,Wai Laing Lee, Lei Wang, Hong Zheng and Chung-Kai Chow and entitled“SYSTEM AND TECHNIQUES FOR SEISMIC DATA ACQUISITION.”

The invention disclosed herein is related to application Ser. No.09/153,864, filed Sep. 16, 1998, by inventors Joel Page, Edwin De Angel,Wai Laing Lee, Lei Wang, Hong Zheng and Chung-Kai Chow and entitled“POWER ON RESET TECHNIQUES FOR AN INTEGRATED CIRCUIT CHIP.”

The invention disclosed herein is related to application Ser. No.09/154,241, filed Sep. 16, 1998, by inventors Joel Page, Edwin De Angel,Wai Laing Lee, Lei Wang, Hong Zheng and Chung-Kai Chow and entitled“NOISE MANAGEMENT USING A SWITCHED CONVERTER.”

The invention disclosed herein is related to application Ser. No.09/153,868, filed Sep. 16, 1998, by inventors Joel Page, Edwin De Angel,Wai Laing Lee, Lei Wang, Hong Zheng and Chung-Kai Chow and entitled“CORRECT CARRY BIT GENERATION.”

BACKGROUND OF THE INVENTION

1. Field of the Invention

The invention is directed to digital filtering and, more particularly,to implementations of linear phase FIR sinc filters.

2. Description of Related Art

Systems for conducting seismic exploration are well known in the art. Onland, a plurality of transducers are deployed over a region andconfigured to receive reflections of acoustic signals from differentgeophysical layers beneath the surface of the earth. Seismic sensors areconnected over cables to signal conditioning, digitization and digitalrecording equipment. When utilizing a seismic system, a strong acousticsignal is generated by, for example, setting off an explosion or byutilizing an acoustic signal generator having a relatively high poweroutput. Reflections of the acoustic signals from the geophysical layersare then received at the seismic sensors deployed over a given area andthe signals recorded, typically, for later analysis.

One problem with seismic exploration is that it frequently occurs inremote areas. Once sensors are deployed over a large area and seismicdata gathered, great expense would be incurred if data were corrupted bymalfunctioning sensors or electronics and a seismic survey crew neededto return again to the site, set up equipment and re-gather the data.

Seismic exploration has exacting requirements for seismic sensors andfor the electronics which processes the signals derived from seismicsensors. There is therefore a need to be able to test both the sensorsand related equipment to ensure that both the devices and the associatedelectronics are functioning properly. It is important that the seismicdata gathering equipment be able to synchronize the data gathered withthe explosion used for a measurement. This is somewhat difficult whenthe timing of the explosion with respect to the triggering signal isunpredictable, as it is with, for example, dynamite.

SUMMARY OF THE INVENTION

The invention is directed to a linear phase FIR sinc filter whichimplements filtering for a plurality of channels using multiplexing andwhich spreads calculations across all machine cycles in a way to reducepeak calculation demands.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a network used to collect data from aplurality of seismic sensors in accordance with the invention.

FIG. 2 is a block diagram showing interconnection of a plurality ofremote sensing units in a network configuration permitting high datareliability.

FIG. 3A is a diagram showing the transmission format utilized on thecommand link shown in FIG. 2.

FIG. 3B is a diagram showing the transmission format on the data linksshown in FIG. 2.

FIG. 3C is a diagram showing an exemplary arrangement of a command frameformat in accordance with the invention.

FIG. 3D is a diagram showing an exemplary data frame format utilized inaccordance with the invention.

FIG. 4 is a diagram showing how round trip delay time is measured for aremote station unit.

FIG. 5 is a diagram showing data shift resulting from round trip delay.

FIG. 6A is an illustration used for explaining network synchronization.

FIG. 6B shows synchronization sequences and how network synchronization

FIG. 7 is a block diagram showing chip pin connections and functionalblocks of a RSU shown in FIG. 1.

FIG. 8 is a block diagram showing signal processing of a seismic sensoroutput at a high level.

FIG. 9 is a block diagram showing a prior art approach to implementingthe processing shown in FIG. 8.

FIG. 10 is a block diagram showing an improved approach to seismicprocessing utilizing a polyphase filter in accordance with theinvention.

FIG. 11 shows an improved version of the polyphase filter utilizingcascaded polyphase filters.

FIG. 12 is a graph showing the response of two members of a set ofpolyphase filters.

FIGS. 13-1A through 13-1C, FIGS. 13-2A through 13-2C and 13-3A through13-3C show relative coefficients, response and transform representationsof response of first order, second order and third order sinc filters,respectively.

FIG. 14 is a block diagram showing a linear phase FIR sinc filterimplementation with selectably variable decimations factors.

FIG. 15 is a diagram illustrating the principles of operation of sincfilter number 1 shown in FIG. 14.

FIG. 16 is a block diagram showing functionally how the data illustratedin FIG. 15 are processed in an exemplary implementation.

FIGS. 17A and 17B together illustrate hardware preferably utilized toimplement the sinc filter Sinc#1 shown in FIG. 14.

FIG. 18A symbolically illustrates the operations of shifting andaddition utilized in carrying out implementation of sinc filters sinc#2shown in FIG. 14.

FIGS. 18B-1 through 18B-4 show the mathematics for a similarimplementation for each of sinc filters sinc#3 through sinc#5.

FIG. 19A is a block diagram of a single-control, multiple datapatharchitecture utilized in implementing sinc filters sinc#2 through sinc#5of FIG. 14.

FIG. 19B shows programming or logic used in item 1910 of FIG. 19A.

FIG. 20 is a block diagram showing how a linear phase FIR sinc filtercan be improved by decomposition of the calculations into two stages.

FIG. 21A illustrates a factor of eight decimation such as might beutilized in one configuration of the circuitry of FIG. 14.

FIG. 21B shows the calculations required to carry out the factor ofeight decimation shown in FIG. 21A.

FIG. 21C shows an improved allocation of calculations resulting from thedecomposition of FIR processing into two stages as discussed inconjunction with FIG. 20.

FIG. 21D shows a further improvement in processing allocation resultingfrom equalization of calculation across sampling instances.

FIG. 22 shows a switched power converter of a type known in the priorart.

FIG. 23 shows an improved switched power converter in accordance withthe invention.

FIG. 24 is a schematic diagram of an exemplary break before make circuitof FIG. 23.

FIG. 25 is a timing diagram showing a protocol suitable for use duringpower on reset when using a switched converter power source.

FIG. 26 is a timing diagram showing a protocol suitable for user duringpower on reset when using a regulator power source.

FIG. 27 is a flow chart of a process used during power on reset of apower source.

FIG. 28 shows a plurality of time lines showing clock alignmentassociated with on-chip generation of clocks in accordance with theinvention.

FIG. 29 is a flow chart of a process for programming clocks inaccordance with the invention.

FIG. 30 is a mathematical relationship showing how a multiply and addoperation using rounding is implemented.

FIG. 31 illustrates how the equation of FIG. 30 would be implemented, inblock form.

FIG. 32 is a block diagram showing the logic of how the multiply and addresult of FIG. 31 is utilized for proper care detection.

FIG. 33 is a logic diagram showing the implementation of carry detectcircuit 3240 shown in block form in FIG. 32.

DESCRIPTION OF THE INVENTION

FIG. 1 is a block diagram of a network used to collect data from aplurality of seismic sensors in accordance with the invention. Aplurality of seismic sensors 100 are distributed over a large area. Eachseismic sensor connects to a respective analog to digital converter(ADC) interface 110. The ADC interface 110 converts the analog output ofits seismic sensor into a digital stream for application to a networkinterface referred to herein as a RSU (RSU) 120. The ADC interface can,of course, be designed to accommodate more than one RSU. RSU 120 is,preferably, an integrated circuit chip designed for low powerconsumption and shown more particularly in FIG. 7. RSUs 120 areconnected to a digital telemetry cable 130 as shown more in detail inFIG. 2. A slave line control unit (SLCU) 140 interfaces digitaltelemetry cables 130 to a 32 Mbps line 150. The SLCU is similar to RSU120 except configured to operate in a master mode. SLCU 140 sendsinformation from a digital telemetry cable(s) which it services to thecentral processing and recording unit and passes information from thecentral processing and recording unit to the RSUs on the digitaltelemetry cable 130. The central processing and recording unit 160collects the data from the sensors for geophysical analysis in a mannerknown in the art.

FIG. 2 is a block diagram showing a preferred interconnection of aplurality of remote sensing units (RSUs) in a network configurationpermitting high data reliability. Other network configurations are, ofcourse, possible. The figure shows a plurality of redundant linescomprising digital telemetry cable 130. A command line or command link200 connects to each of the RSUs as described more hereinafter. Each RSU120 connects to each adjacent neighbor over links such as 210 and 220shown in FIG. 2. The remote sensing unit also connects to each nextadjacent neighbor over links such as links 230 and 240 shown in FIG. 2.In a preferred embodiment, the remote sensing unit has 4 data ports,each bidirectional in nature, and which permits a robustness ofinterconnection ensuring high reliability in the return of data from theseismic sensors over the digital telemetry cable 130. The particulardata ports at the remote sensing unit 120 utilized for the data linkreturn of information from the seismic sensors to the central processingand recording unit can be specified by the central processing andrecording unit 160 as described more hereinafter.

The central processing and recording unit 160 sends commands toindividual RSUs (RSU), groups of RSUs or to all RSUs over the commandline. The command line utilizes two wire differential Manchesterencoding and each RSU utilizes a phase lock loop to effectuate clockrecovery from the incoming command line data. In a preferredimplementation, the PLL clock recovery locks at a clock rate 16 timesthe line rate of the command line.

The network shown in FIG. 1 operates normally in a poll-select mode. Thecentral processing and recording unit operates as a network masterstation which continuously polls one or more RSUs on an ongoing basis.When a station is polled and has information to send to the central, itreturns a flag or a flag and data indicating that data is to be sent oris sent concurrently. When the RSU is selected for data transmission(i.e. authorized to transmit data by the central), the RSU sends databack over the data link. The particular port utilized to send the datahas been previously set by the central processing and recording unit byinformation transmitted over the command line. Thus, the centralprocessing and recording unit controls the individual ports utilized ineach RSU and thus defines the data ports in use at each RSU for thereturn data link.

FIG. 3A is a diagram showing the transmission format utilized on thecommand link shown in FIG. 2. The central processing and recording unit160 sends a continuous stream of command frames over the command link asillustrated in FIG. 3.

Due to the nature of the data link, the slave nodes have always accessto the data link. Setting of the slave nodes into a transmit or into arepeat mode on the data link is controlled by the master node. Themaster is usually only listening to the data link. The slave nodestransmit seismic data frames, status frames and auxiliary frames to themaster node on the data link.

FIG. 3B is a diagram showing the transmission format on the data linksshown in FIG. 2. A plurality of data frames 400 are transmittedrepeatedly on the data link. Each data frame is separated from anadjacent data frame by zero or more idle slots. The actual number ofidle slots employed between data frames is determined by the distancebetween nodes. The number of idle slots is utilized to ensure that therewill be no collisions due to propagation delays on the link.

The data link may be operated selectively in a high rate mode and in alow rate mode. The RSU may operate in a number of operational modes. Ina booting mode, a number of data links and command link transmissionparameters are determined (e.g bit synchronization, framesynchronization), node configuration, etc. Every node/channel isassigned a logical network address during the booting mode.

In an initialization mode, the application modules in the RSU will beprogrammed through the telemetry interface (TMI). This involvesdownloading of control register values, the setting of program andcoefficients for the digital signal processor and auxiliary nodes.

In an acquisition mode, a continuous poll/configuration/NOP command bitstream is received and a seismic/status/auxiliary word bit stream istransmitted to the slave line control unit 140 for passing to thecentral processing and recording unit 160.

The RSU can be set in a command loop back mode which is used for themeasurement of node distances from the central. In the loop back mode,the received command bit stream will be looped back and transmitted onthe data link back to the central. This can be optionally done withscrambling and descrambling to achieve desired spectral characteristics.

In a diagnostic node, the RSU can be utilized for detection of datalinks having degraded bit error rate performance. In this mode, the lastnode on a line is programmed to transmit a downloaded diagnostic patterncontinuously and all other nodes detect the occurrence of the diagnostic(unique) pattern in the repeated bit stream. This, too, may beselectively scrambled.

Each RSU may operate in an SPI master mode in which it serves as amaster node for a serial peripheral interface (SPI) bus. Alternatively,the RSU may operate in a SPI slave mode.

In a test mode, the internal telemetry functionality will be verified byrunning a test procedure from the central processing and recording unitoperating as a system network controller.

FIG. 3C is a diagram showing an exemplary arrangement of a command frameformat in accordance with the invention. The command frame formatutilized on the command link begins with a frame sync pattern 500. Apoll command 510 and a poll address 520 are utilized to specify the typeof poll and the address of the station(s) designated to respond. Theconfiguration address 530 and configuration command 540 together withparameters 550 are utilized to set configuration at one or more RSUs.The TSG data 560 is utilized to send information for driving a testsignal generator in the RSU. The frame ends with a frame check sequence570, preferably using a cyclic redundancy code (CRC) check sum.

FIG. 3D is a diagram showing an exemplary data frame format utilized inaccordance with the invention. The seismic data frame has a fixed lengthof 448 bits, configured as follows: The frame begins with a scramblinginitiation pattern 600. It is followed by a frame sync pattern 610indicating the start of data. The source address 620 identifies the RSUand, if more than one channel is utilized on the RSU, the channel whichis the source of the data. A particular type of data frame can bespecified in fields 630. A time tag 640 permits certain timingadjustments to be made. A plurality of seismic samples 650 then follow.Certain status flags can be sent in field 660. The seismic data frameends with a CRC frame check sequence 670.

On the command link, frame synchronization is based on transmission ofan eight bit long frame sync bit pattern in every transmitted commandframe. The sync pattern alternates between a pattern A and a pattern Bin consecutive command frames. Pattern A is the inverse of pattern B. Arest command occurs after command number 73 in the polling sequence andcontains two C patterns which are used for detection of the remainder inthe polling period.

There is no separate frame synchronization procedure for the data linktransmission in the RSUs. The data link transmission are phase locked tothe command link transmission.

The addresses for the individual RSUs are assigned by the master unit asa function of distance and polling occurs in address sequence, beginningwith the closest RSU.

FIG. 4 is a diagram showing how round trip delay time is measured for aremote station unit. There are two major adjustments used insynchronizing the network. One adjusts for round trip delay. The otheradjusts the timing of data gathering.

When adjusting for round trip delay, the central stations 400 places aparticular RSU into a loop-back mode and sends a bit pattern, such as0110100 over the command link. In an exemplary embodiment, the data linkis operated at 4 times the rate of the command link. Since the clocksare synchronized, one bit from the command link will be sampled fourtimes for transmission over the data link.

FIG. 5 is a diagram showing data shift resulting from round trip delay.In the example shown in FIG. 5, the loop backed version of the sampledsynchronization pattern is compared in phase with the expected returnsignal. In the example shown in FIG. 5, two clock units of delay areexperienced during the round trip. A single unit of delay added to thepath will be traversed twice, once in the outgoing and once in thereturn direction, thus equalizing the delay to what it should be.

FIG. 6A is an illustration used for explaining network synchronization.A free running counter 600 runs at an exemplary 4 MHz rate. It is resetupon the first CCA or CCB pattern which occurs after a SYNC signal. Thelatch contains the selected CCA or CCB pattern. If a SYNC signal isn'treceived, nothing happens. When the next SYNC signal is actuallyreceived, the counter is reset and the amount of any error can bedetermined. These relationships are illustrated in FIG. 6B.

FIG. 7 is a block diagram showing chip pin connections and functionalblocks of an RSU shown in FIG. 1. As shown in FIG. 7, the command linkreceiver 715 connects to and receives commands over the command link200. A set of buffered outputs are available for external use. Thecommand link receiver passes commands to command decoder 720 where thecommands are decoded or interpreted and appropriate commands and datasent over bus 700 to the various connected devices shown in FIG. 7 asfunctional blocks connected to the bus.

This chip shown in FIG. 7 also includes a separate digital signalprocessor (DSP) data bus 705. This bus is utilized in connection withthe processing of signals received from ADC interface 110 over inputsMDATA[1], MDATA[2] and MDATA[3]. Certain portions of the data filteringdiscussed hereinafter occur in modulator data interface 730 with theremainder executed in the digital signal processor 735. The allocationdescribed hereinafter is preferred, but other allocations are possible.When the processing of the incoming digital signals is completed by theModulator Bus Interface and the DSP and it is desired to transmit thedata to the central processing and recording unit 160, the data isapplied through data FIFO 740 to data transceiver 745. The datatransceivers 745 include four ports referred to generally as DATAA,DATAB, DATAC and DATAD in FIG. 7. Those four ports are utilized toachieve the network conductivity described in conjunction with FIG. 2.

General purpose I/O (GPIO) 750 can be used to pass signals to one ormore attached devices, such as passing control signals to ADC interface110. The serial peripheral interface 755 can likewise be utilized tocommunicate with external peripherals and, in one application, can beutilized to upload code to programmable devices on the ADC interface110.

The regulator/SC converter 770 is utilized to provide a programmableDC-DC converter to permit selective voltage levels to be generated forthe chip. This is discussed more hereinafter.

The TSG buffer and filter 760 is utilized to send test signal data tothe ADC interface 110 for testing purposes.

The scratch pad memory 780 is utilized for calculations on an as neededbasis. The watch dog timer 790 ensures that the DSP data bus 705 doesnot hang up without being noticed.

As part of the bootup/initialization of the network, the centralprocessing and recording unit 160 broadcasts a rough delay value to allRSUs. That value is the same for all RSUs and is stored in a registerwithin the chip 120 for delay equalization purposes. After that is done,the central processing and recording unit 160 polls each of theindividual RSUs, one at a time, sends a loop back command to the RSU tocause the data received over the command link to be looped back over oneof the data links to the central processing and recording unit 160, thuspermitting the central processing and recording unit 160 to measure theround trip delay from the central to the RSU and back. Once the amountof delay is determined based on the round trip delay, the centralprocessing and recording unit 160 will load a register of the individualRSU with a fine delay value to be used for correcting for differences indelay. The amount of fine adjustment loaded in each RSU is different andis based on the described measurement of the round trip delay time. Thegoal is to have all nodes sampling at the same point in absolute time sothat data received at the central processing and recording unit fromeach of the nodes will have the same time base.

FIG. 8 is a block diagram showing at a high level signal processing of aseismic sensor output. The analog signal from seismic sensors 100 ispassed through ADC interface 110 to certain decimation filteringimplemented on RSU 120 as described more hereinafter and then through tothe central processing and recording unit 160. In a preferredembodiment, it is received from the ADC interface 110 as 512 kHz, 1 bitdelta-sigma data. The decimation on RSU 120 converts the one bitdelta-sigma modulated data into 24 bit sample data having a recurrencerate ranging between 250 Hz and 4 KHz depending upon the settings of thedecimation filter. This filtering will be discussed more hereinafter.

When the arrangement shown in FIG. 8 is utilized, there is a problem.The acoustic source utilized to gather seismic data is not synchronizedwith the seismic data acquisition system clock. This is particularlytrue when dynamite is utilized as the source of the acoustic impulse.Even if the triggering signal for the dynamite is synchronized with theseismic data acquisition system clock, there is an uncertain delay fromthe application of the triggering signal to the actual detonation of thedynamite. As a result, it is necessary to realign all channels of datain the time domain based on the actual detonation point. For a 512 kHz1-bit sample rate, the decimated output data rate is only 1.0 kHz, butthe time resolution of synchronization is required to be 4.0microseconds or less. There are a number of sources of delay from theshooting time to the time of receiving data from all channels. The delayincludes the network propagation delay, discussed above, and filtercalculation delay.

FIG. 9 is a block diagram showing a prior art approach to solving theproblem discussed in conjunction with FIG. 8. In the prior art, toachieve that synchronization, the one bit signal from the ADC interface110 was applied to a data RAM buffer 900 and stored there until asynchronization signal was received from control logic 910 indicatingthat the shot had occurred. The data samples were then read beginningwith a point in the data RAM buffer which corresponded to the neededamount of delay to synchronize the data with the shot. Once that pointwas identified, data was passed to a digital processing chip. Therevariable decimation filtering would occur resulting in an N-bit 1.0 kHzoutput signal.

The approach shown in FIG. 9 has several disadvantages. First a longsystematic delay requires a large amount of storage, so much so that anadditional RAM chip is required before decimation in order to store thedata after the shot at the resolution of the sampling rate. Thatincreases expense and reduces reliability. There is also a need forextra control logic. For example, at a 512 kHz sample rate, for eachdata conversion channel, a systematic delay of 50 milliseconds (typical)needs a RAM size of 25.6 kilobits. If the chip shown in FIG. 7 handlesthree data conversion channels as the chip shown in FIG. 7 does, itwould require 76.8 kilobits of storage.

FIG. 10 is a block diagram showing an improved approach to seismicprocessing utilizing a polyphase filter in accordance with theinvention. After decimation filtering 920, a polyphase all-pass linearphase FIR filter is implemented and does the selective phase adjustmentneeded to bring the data into alignment with a shot. In this case, theall pass linear phase FIR filter adds a group delay of (N−1)*4.0microseconds. By storing and selecting a number of filter parametersets, N different all-pass filters can be selectively implementedresulting in a polyphase filter or phase shifter. Each set ofcoefficients provides a group delay of i*4.0 microseconds, wherei=0,1,2, . . . , N−1.

If the output rate is 1.024 KHz and the synchronization resolutionrequired is 4.0 microseconds, then one could implement selective delaysbetween 0 and 50 msec at 4 μsec resolution by using a group polyphasefilter with 256 sets of coefficients. The particular set of coefficientsselected to add a group delay to the output data depending on the timeof occurrence of the shot. Thus, each set of filter coefficients canimplement a phase shifter having a discrete group delay of i*4 μsec,where i=0,1,2, . . . , 255.

When the central processing and recording unit 160 detects a shot, itsends a command (e.g. broadcast) specifying a time value for the shot.The time value can be established, for example, by detecting theexplosion a the central processing and recording unit or by adding aknown delay from the triggering instant. Upon receipt of that command,the amount of shift required to adjust the phase of the sampling to thetiming of the shot is determined and a filter coefficient set isselected to impart the appropriate group delay to the polyphase all-passlinear phase FIR filter 1000. The polyphase filter thus makes the timingadjustment needed to synchronize with the shot. Thus, the phaseadjustment imposed by the polyphase all-pass linear phase FIR filter1000 varies from shot to shot and ensures that the data is synchronizedwith the shot. Further, since the decimation filtering process 920removes the HF noise and lowers the data rate, very little storage isrequired.

In an exemplary implementation, a 256:1 decimation filter can beutilized with a sampling frequency Fs of 256 kHz with N_(tot) taps. Thecoefficients of the filter can be decimated by the ratio 256 by pickingup coefficients every 256 points. The coefficient of one set ofpolyphase filter is formed and the number of its taps is N_(tot)/256.There are thus totally 256 different sets of N_(tot)/256-tap linearphase FIR filters obtained from the decimation filter, each having adata rate equaling 1.0 kHz. Each set has a group delay difference of 4.0microseconds from its adjacent sets of filter coefficients. Thus a phaseshifter can be described as h_(p(ij))=h_((j−1))*256+i, where i equals aninteger from 1 to 256 and represents a number of the set and where j isa number from 1 to N_(tot)/256 which represents the numbering of thecoefficients.

The coefficients for the ith set of coefficients for a polyphase filterare inversely symmetrical to the (256-i)th set of coefficients. Thus,the storage required to store the coefficients for the polyphase filtercan be reduced by a factor of 2 by taking advantage of that symmetry.

FIG. 11 shows an improved version of the polyphase filter which utilizescascaded polyphase filters. Several benefits can be achieved fromsplitting a polyphase filter into two polyphase filters. First thecalculations needed for the cascade filter is about the same as thesingle stage polyphase filter but a reduced number of taps is required.In the example discussed in conjunction with FIG. 11, the polyphasefilter 1 utilizes 16 sets of coefficients, each one differing from anadjacent set of coefficients by 64 μsec. Polyphase filter 2 thenprovides for 4 μsec resolution within the 64 μsec windows provided bypolyphase filter 1. Thus, only 32 sets of coefficients are required inorder to specify the 256 4 μsec windows required to achieve theresolution needed to synchronize with the shot over a 50 msec interval.If a single stage polyphase filter were utilized, then 256 sets ofcoefficients would be required. Thus, the coefficient storagerequirements for the polyphase filter are reduced considerably bydividing the polyphase filter into two polyphase filters. Also, each setof cascade polyphase filter coefficients is shorter than a set of singlestage polyphase filter. Even if a cascade calculation of two filters isneeded, the total calculation amount is about the same as needed in thesingle stage polyphase filter.

Additionally, using a 2 stage polyphase filter, there is an ease ofaddressing associated with the selection of the overall delay requiredfor synchronization to the shot. The amount of delay can be specified asa single byte with the 4 most significant bits specifying which of the64 microsecond windows should be established by polyphase filter 1 andthe least significant bits specifying the 4 microsecond window withinthe 64 microsecond window required to synchronize with the shot. Thus, asingle word can be utilized to select the coefficients for bothpolyphase filter 1 and polyphase filter 2.

FIG. 12 is a graph showing the response of two members of a set ofpolyphase filters. FIG. 12 shows two curves reflecting the response of apolyphase filter, each curve representing the response for a respectiveset of coefficients. In essence, the response is substantially identicalbut shifted in phase by a fraction of a sampling interval.

The polyphase filter described herein is much better than prior arttechniques because the polyphase filter can be implemented on thedigital chip resulting in the elimination of the extra RAM chip and itscorresponding cost and reliability problems. It is suitable for use inany case where real-time high resolution synchronization is required andit reduces ROM and calculation power needed over that required by theprior art.

In the chip architecture shown in FIG. 7, the polyphase filter andlinear phase FIR filter and a IIR filter are implemented using thedigital signal processor 735.

An exemplary set of coefficients for polyphase filter 1 is set forth inAppendix A. An exemplary set of coefficients for polyphase filter 2 isset forth in Appendix B.

FIGS. 13-1A through 13-1C, FIGS. 13-2A through 13-2C and 13-3A through13-3C show sample weighing (coefficient values, response and transformrepresentations of response of first order, second order and third ordersinc filters respectively.

The decimation filtering 920 shown in FIG. 10 includes a sinc filterwhich receives the output of the ADC conversion accomplished by ADCinterface 110. The sinc filters of the prior art consume more power thanwas desirable for the low power implementation of the invention. Sincdecimation filters are preferably used because they have well behavedtransfer functions and high attenuation at the alias frequencies. In thetime domain, they have relatively few taps and use small integercoefficients. A sinc filter can be realized in at least two ways. In oneform, a sinc filter can be expressed as a cascade integrate-comb (CIC)filter. Such a filter has the following transfer function:$\begin{matrix}{{H(Z)} = \frac{( {1 - Z^{- R}} )^{N}}{( {1 - Z^{- 1}} )^{N}}} & ( {{Equation}\quad 1} )\end{matrix}$

where R is the decimation ratio and N is the order of the filter. Thiscan be realized as a combination of integrators and differentators.

Alternatively, a sinc filter can be expressed as a linear phase FIRfilter. In this case:

y(n)=h ₀ ·x(n)+h ₁ ·x(n−1)+ . . . +h _(m−1) ·x(n−m+1)  (Equation 2)

where M is equal to the number of taps and where the taps are symmetric.

A CIC sinc filter implementation can be constructed of integrators anddifferentators in either a direct or cascade structure. While the CICimplementation uses only additions and permits easy achievement ofvariable decimation ratios, it uses considerable power and is thereforenot suitable for low power filter design. In addition, the accumulatorlength grows very fast with filter order and decimation ratio which inturn also increases power consumption.

An linear phase FIR sinc filter implementation, on the other hand, hasmore complicated hardware requirements and more complicated operatingsequences and would not likely normally be chosen for an IC design, but,in this implementation, it has the advantage that power savings can beachieved since (1) the quantities of computation required are decreased,(2) the register length can be kept at 24 bits or less, (3) one bitinputs permit table lookup of coefficients, (4) the coefficients aresmall and integer and (5) the filter can be implemented with shifts andadditions.

FIG. 14 is a block diagram showing a linear phase FIR sinc filterimplementation with variable decimation factors. Variable orderdecimation in accordance with the invention can be achieved by switchingin or out, selectively, a plurality of sinc decimation filters. A twostage decimation process is illustrated. The first stage, in a preferredembodiment, includes a fifth order, 36 tap Linear phase FIR sinc filterused to decimate a 1 bit 512 kHz input by a factor of 8 to a 64 kHz 17bit input. The output of the first stage sinc filter is applied to apipe line arrangement of sinc filters which can be selectively activatedin sequence to achieve desired decimation ratios. In the examples shown,decimation ratios of 16, 32, 64, 96 and 128 can be selected. Otherarrangements can be implemented to achieve different ratios as desired.The sinc 1 linear phase FIR filter implementation has the advantage thatit can be implemented with lookup tables and additions (see equation 2).The tables are small enough for direct implementation because the filtercoefficients are symmetrical and because partial results areanti-symmetric for one bit inputs. Using these symmetries, one canreduce the ROM size required to about 25% of what would otherwise havebeen required.

FIG. 15 is a diagram illustrating the principles of operation of sincfilter sinc#1 shown in FIG. 14. The 512 kHz one bit input to the sinc 1first stage input is fed into a serial register 1500. There is a centralline 1510 which forms an axis of symmetry for analysis purposes. Oneither side of the symmetry line, 8 bit words are defined, namelyword#1, word#2, word#3 and word#4 as shown in the figure. In thisimplementation, the register is 36 bits long. As a result, two bits,namely X₀ and X⁻¹ are left over on the left edge of the register. Thesebits will be referred to as the “head” bits. In addition, two bits areleft over at the right extreme of the register, namely bits X⁻³⁴ andX⁻³⁵. These two bits are referred to as the “tail” bits. When multipliedby respective coefficients H_(i) each of the bits in the register forman output.

FIG. 16 is a block diagram showing how the data discussed in conjunctionwith FIG. 15 are processed in an exemplary implementation. In FIG. 16, aconvenient way of multiplying the incoming bits by the coefficients ofthe sinc filter is shown. A plurality of lookup tables 1600, 1610 and1620 (implemented either as ROM or logic) are utilized for determiningthe corresponding output value for various combinations of bit values inthe word used to access the look up table. The output value relates tothe multiplication of those bits by the coefficients. As a first step,the head and the tail of the 36-bit data structure discussed in theprevious figure are combined in respective head and tail registers andutilized to access the look up table or equivalent logic to produce anoutgoing value Y0. In step 2, word#1 is utilized to look up acorresponding value Y1 in ROM 1600. In step 3, word#2 is utilized tolookup a value Y2 from ROM 1610. In step 4, word#3 is “twisted,” meaningthe bit order is reversed, and utilized to look up the value Y3 in ROM1610. In step 5, word#4 is twisted and utilized to look up the value Y4from 1600. The values Y0, Y1, Y2, Y3 and Y4 are summed to produce theoutput. The use of lookup tables in this manner reduces the amount ofcalculation required and thus power consumption.

Although the calculation process has been described here at a functionallevel, the actual circuitry utilized for implementation is describedmore in conjunction with the following figures.

FIGS. 17A and 17B together illustrate hardware preferably utilized toimplement the sinc filter number 1 shown in FIG. 14. Returningmomentarily to the modulator data interface 730 of FIG. 7, the threedata inputs MDATA (1), MDATA (2) and MDATA (3) are applied to themodulator data interface. These inputs correspond to the channel 1(CH1), channel 2 (CH2) and channel 3 (CH3) inputs to respective buffers1700 (e.g., 1700-1, 1700-2, 1700-3). Words stored in buffers 1700 aretransferred to respective pages of RAM 1710. The head and tail valuesare written to respective head and tail registers 1720 and 1730. Thecombined values from the head and tail registers of a given data planeare combined to form a small look up table address, which in the exampleshown, is a ROM address which is utilized as shown in FIG. 17B.Similarly, the words stored in a particular data plane 1710 are read outand passed to a large look up table (a ROM in the example illustrated)in either regular or twisted form to facilitate the lookup. Twisting ofthe word is accomplished in a twist multiplexer 1740 which passes dataeither in regular or bit reversed order to the output depending on thevalue of the twist control input. Control logic 1750 provides controlsignals to portions of the chip shown in FIG. 7 and to the second stagesinc filters. A sync signal is received which specifies time zero forpurposes of establishing sample intervals. Thus, the reading and writingof data will be based on the same sample intervals as the remainder ofthe chip. A three channel handshake is utilized to indicate a requesthas been received (data ready) and to receive back an acknowledgement(when no error occurs). A head select line permits early storage of thehead portion of the register bits so that it will be available whenneeded in processing. A small ROM address and the ROM address from FIG.17A are applied respectively to small ROM 1760 and large ROM 1770 ofFIG. 17B. The lookup table output values are selectively applied to anadder via switch multiplexer 1775 which selects the input value to bepassed to adder 1780 in accordance with incoming control signals. Theoutput of 1780 is fed back to the input via an accumulator 1790. In thismanner, the outputs y0, y1, y2, y3 and y4 as discussed in FIG. 16 arecombined and passed as a 17-bit output to a second stage sinc filter at64 KHz rate.

The second stage sinc filters include sinc#2, sinc#3 (1), sinc#3 (2),sinc#4 and sinc#5. The mathematics for expressing each of these filtersis set forth in FIGS. 18A and 18B. Each of those sinc filters isimplemented using a number of words and a number of additions.

FIG. 18A symbolically illustrates the operations of shifting an additionutilized in carrying out implementation of sinc filters #2-#5 shown inFIG. 14. In the drawing, each binary bit x_(i) is multiplied by acoefficient which is a power of 2. Multiplication by a power of 2 isequivalent to a shift by a number of places equal to the exponent of thepower. When a coefficient has a value which cannot be expressed as aneven power of 2, it is decomposed into two terms which when summedtogether result in the appropriate value for that term. As shown in FIG.18A, for sync 2, the third term has a coefficient of 6, which is not aneven power of 2. However, as shown in the dashed box in the right handpart of the equation for sinc#2, a coefficient of 6 can be stated as4*X⁻²+2*X⁻². This term is thus equivalent to 6*X⁻².

FIG. 18B shows the expressions which can be used to implement sincfilters sinc#3-sinc#5.

FIG. 19A is a block diagram of a single-control, multiple datapatharchitecture utilized in implementing sinc filters sinc#2 through sinc#5of FIG. 14. The shifting and the additions necessary to implement aparticular sinc filter as discussed in conjunction with FIGS. 18A and18B are implemented in the circuitries shown in FIG. 19. A sequencecontroller 1900 receives the handshaking from the first stage aspreviously discussed, a signal indicating whether one or three channelsare implemented, clock rate to be used and a decimation factor. Aplurality of commands are read from the command table such as ROM 1910and the commands sequentially read out are applied to the commandexecution unit 1920. The 16 kHz 17-bit signals from the first stagecomprising a 16 bit value and a sign bit are applied to respective dataplanes 1930-i which act as incoming buffers. As the respective wordsemerge from the buffer, they are stored in respective individual pagesof RAM 1940. As individual words are read out of individual data planes1940, they are applied to shift multiplexer 1950 where they areselectively shifted in accordance with the shift control code applied tothe mux and applied to one input of adder 1960. As before, the output ofthe adder is applied to the input of an accumulator 1970 and that outputis applied to a second input of the adder. The output of the adder caneither be recirculated over gate 1980 or applied as a 24-bit output tothe digital signal processor over mux 1990. By controlling the sequenceof the data circulation, in a pipeline arrangement, one can implementthe multiple sinc filters needed for a particular decimation ratio.Thus, the second stage shown in FIG. 14 can be implemented using thearrangement shown in FIG. 19.

Advantages of the single-control multiple datapath are:

A. Gate clocks to each datapath independently allows unused channels tobe turned “off” for low power.

B. Run the complete block at a lower clock rate than for a design wherea single datapath is used for multiple channels. This provides a linearreduction in clocks. (i.e. if 3 channels on 1 datapath require 1 MHZ,then 3 channels on 3 datapaths can be done in 1 MHZ.3=333.3 kHz.)

C. Arbitrarily add or remove channels to the design very easily with nomodification to the control.

D. All channels generally are guaranteed to run the same code, sowriting the code is easier (only consider 1 channel, not 3), and themultiple channels don't need to be interleaved in time (i.e. don't needto split code for ch 1, ch 2, ch 3 and so on).

E. The code for each channel must still be interleaved with the incomingdata to spread out the computations so that the minimum clock frequencycan be used.

FIG. 19B shows programming or logic used in item 1910 of FIG. 19A. Theexample shown in FIG. 19B follows the ordering needed to implement theA-O mode multiplexing discussed hereinafter in conjunction with FIG.21C. If implemented in logic, there is a main routine, each activated byone of eight command lines. The main routine calls subroutines, in thiscase, also implemented in logic. In the example shown in line 1 of themain routine, there are two subroutine calls, the first to sinc 3(1)_(A)and the second to sinc 5 _(O). Each of those routines is implemented inthe subroutine logic or an equivalent RAM. The subroutine sinc 3(1)_(A)comprises two lines of microcode implemented in logic and the subroutinesinc 5 _(O) comprises six lines of microcode implemented in logic.

FIG. 20 is a block diagram showing how a linear phase FIR sinc filtercan be improved by decomposition of the calculations into two stages. Itis possible to reduce the hardware requirements and the calculation rateneeded for implementing a particular sinc filter by splitting theprocessing across two stages. This principle is illustrated in FIG. 20in which a data value is multiplied by a respective set of coefficientsand their values delayed and summed with subsequent products. If theprocess shown at the top half of FIG. 20 where to be separated into twophases, namely first an accumulate phase (A phase) and then an outputphase (O phase), as shown in the bottom half of FIG. 20 the total numberof registers needed can be reduced from 4 to 2 resulting in considerablepower savings and in savings of silicon real estate.

FIG. 21A illustrates this principle using a factor of eight decimationsuch as might be utilized in one configuration of the circuitry of FIG.14. The pipeline shown in FIG. 21A will be used as an example comparingthe calculation requirements at various points in time using thetechniques described herein.

FIG. 21B shows the calculations required to carry out the factor ofeight decimation shown in FIG. 21A. One can see that various amounts ofcalculation occur at alternative sample instances when no multiplexingis employed. That is, calculations are fairly intensive at one instantbut non-existent at another instant. Even during those instances inwhich calculation occurs, the amount of calculation varies from sampleinstant to sample instant. The clock rate must be high enough to handlethe largest number of calculations per sample insert.

FIG. 21C shows an improved allocation of calculations resulting from thedecomposition of FIR processing into two stages as discussed inconjunction with FIG. 20. Using the A-O mode of multiplexing describedin conjunction with FIG. 20, the amount of calculation is spread outover all instances but the peak amount of calculation required isconsiderably reduced. Since the peak amount of calculation is reduced,the clock rate can be reduced, saving power.

FIG. 21D shows a further improvement in processing allocation resultingfrom equalization of calculation across all sampling instances. Here,each sample instant has an identical amount of calculation going on. Thearchitecture of the second stage sinc filter as shown in FIGS. 19A-19C,permits each of these options to be implemented as desired. Because ofthe flexibility of that architecture, any of the approaches shown inFIG. 21B, 21C or 21D can be carried out.

It one were to estimate the calculations required for the different sincfilter approaches shown in FIGS. 21B, 21C and 21D, assuming that anequivalent computation rate was equal to the sample frequency times thenumber of additions, times three channels, where one addition means one24-bit addition/subtraction, one would observe the following results.

R = 16 R = 32 R = 64 R = 96 R = 128 CIC 9,792 10,656 10,512 11,61611,592 DIRECT STRUCTURE CIC 10,080 8,034 7,458 5,836 5,673 CASCADESTRUCTURE FIR 1,832 2,216 2,024 2,024 1,928 STRUCTURE

One can see that the inventive linear phase FIR filter structureimplementation described above results in a greatly reduced computationrate when compared with direct or cascade CIC structures. The reducedcalculations will result in significant power savings.

Additional power savings can be achieved through the construction ofregulator/switched converter 770 shown in FIG. 7. Switch converters areknown in the art. One such switch converter is described in an articleentitled “HIGH-EFFICIENCY LOW-VOLTAGE DC-DC CONVERSION FOR PORTABLEAPPLICATIONS” by Anthony J. Stratakos et al. of the University ofCalifornia at Berkley and described at pages 105-110 of the IWLPD '94Workshop Proceedings. FIG. 22 shows a switched power converter of a typedescribed in the article. A square wave input is applied in parallel tothe gates of a PMOS and NMOS device. The PMOS and NMOS devices areconnected in series. An output from the junction of the drain and sourceof the PMOS and NMOS devices is applied to an inductor L1 and the otherend of the inductor is provided to a smoothing capacitor C1 and anoutput line to provide voltage for the integrated circuit chip.

FIG. 23 shows an improved switched power converter in accordance withthe invention. In accordance with the invention, the prior art switchingconvertor is modified by inclusion of a break before make circuit 2300.This ensures that none of the devices is turned on substantiallycompletely before the other device is turned off, thus avoidingswitching problems of the prior art and their accompanying powerconsumption.

The implementation of this break before make circuit 2300 is shown morein detail in FIG. 24. The clocking input is applied to a NAND gate I1and a NOR gate I2. The A input on each gate is inverted. The output ofthe gates I1 and I2 drive respective chains of inverters, the output ofwhich is fed back to one of the inputs of the gates by inverters I7 andI8, respectively. Thus, when enabled, the circuit of FIG. 24 ensuresthat one of the two series transistors of the switched converter isopened (turned off) before the other is closed (turned on).

The circuit shown in FIG. 23 has yet other benefits of that shown in theprior art. As shown in FIG. 23, the square wave generator 2320 whichdrives the break before make circuit 2300 is controlled by a moderegister 2310. The mode register permits the chip voltage to be set bycommands sent over the command link 200 and applied to the regulator/SCconverter over the TMI bus shown in FIG. 7. The value in the moderegister controls both the duty cycle of the square wave, which permitsthe output voltage V_(chip) to be set, as well as the phase of thesquare wave generated. The ability to adjust and control the phase ofthe square wave is particularly critical because the switching generatedby the switched converter has a sharp rise time and fall time whichtranslate into relatively high frequency components which can be coupledeasily as noise into other circuits. By being able to control thephasing of the square wave, the noisy transition instants in theswitching converter can be set to occur at a time when sensitive signalprocessing functions are not going on. For example, during chargetransfer using a switched capacitor input circuit to sample the analogoutput value of a seismic sensor, one would prefer to have as littlenoise as possible in the neighborhood. The switching transition instancefor the switch converter can be set so as to occur when such sensitivecharge sampling operations are not occurring. The power on reset circuitshown in FIG. 7 of the drawings applies to protocol which isadvantageous in ensuring correct startup of the chip.

FIG. 25 shows a set of timing diagrams which describe that operation.When the 5 volt VDD is first applied (2500) it rises from 0 volts to itssupply value of approximately 5.0 volts. Once the value of the appliedVDD rises to a point which exceeds three times the threshold voltage ofthe devices in question, the power on reset circuit is activated (2510)and the phase locked loop begins its oscillation. When power is firstapplied, the duty cycle for the switched converter is held to unity,that is, it is always on. Thus, the output voltage of the switchedconverter rises above its 2.5 volt VDD line and reaches 5.0 volts(2520). After the output of the switched converter is stabilized at 5volts, the duty cycle hold on the switched converter is released and theswitched converter seeks the output value programmed for it by the moderegister (2530) and the switched converter begins to seek its programmedvalue. After a time T_(SC) _(—) _(SETTLE), the 2.5 VDD output, orequivalent value set in the mode register, is stabilized and the holdapplied to all clocks is released and the chip begins to operate.

FIG. 26 shows a similar power on reset operation utilized when the powersource is controlled by a regulator. However, in this case, the switchedconverter is not utilized but rather a regulated version of an externalpower source is used. The external power source functions as the 5 voltVDD line did in the discussion of FIG. 25 and time lines havingcorresponding labels to those shown in FIG. 25 behave as describedpreviously. However, since the switched converter is not utilized, thosetime lines are not shown. In addition, the 2.5 volt VDD line beginsrising gradually as plot power is applied until it reaches a stable, inthis case 2.5 volt level. At that time, after expiration of timeT_(resetz), the hold on all clocks is released and the chip begins tofunction. An optional reset mode is used in a third mode which the timerequired for reset is reduced to a few clock cycles. This is used fortesting on, for example, an industrial IC tester. This is possiblebecause the voltage ramps on such a tester are well defined and a longtime for voltages to stabilize isn't needed.

FIG. 27 is a flow chart of the process described in conjunction withFIG. 25. VDD is applied (2700) and when the applied VDD exceeds 3V_(th)(2710) the PLL starts (2720). The SC duty cycle is set to hold at about100% (2730) and when the SC output nears VDD, the duty cycle hold isreleased and the switched converter is allowed to have settled to itsnominal voltage established in the mode register (2740). Once it issettled, all clocks are released with the next clock reset pulse (2750).

The clock recovery and reset logic 725 shown in FIG. 7, contains a phaselock loop which is phase locked to the command line 1 Mbps Manchesterencoding rate. In Manchester encoding, an up transition or a downtransition in the center of the sample window is interpreted as a logic1 or a logic 0. The PLL locks on to these transitions, although theoutput of the PLL is preferably, in this example, 16 times the 1 Mbpsrate of the Manchester encoding. This 16 Mbps clock signal is utilizedas a master chip clock and all clocks on the chip are derived from thisclock.

It has been found particularly advantageous to generate all clocksinternal to the chip so that they coincide with the rising edge of thechip clock. All noise critical clocks provided external to the chip,such as ones provided to the ADC interface 110 shown in FIG. 1 arecreated on the falling edge of the chip clock.

All clocks on the chip shown in FIG. 7 are programmable. That is, thedivision ratio used to obtain a particular clock rate from the chipclock can be programmed. Not only that, they can be programmed duringthe operation of the chip. The registers setting the dividers for thevarious clocks can be programmed over the TMI bus using informationreceived over the command line. Thus, the central processing andrecording unit 160 can set individual clock rates on the chips. Thearrangement execution of a change in the programming for a particularclock can occur only when a chip sinc pulse occurs. This occurstypically at a 32 kHz rate.

FIG. 28 shows a plurality of time lines showing clock alignmentassociated with on-chip generation of clocks in accordance with theinvention. These time lines illustrate the principles just discussed. InFIG. 28, CLK 16 is the clock to which all other clocks are locked. Aplurality of additional clocks, CLK 8, CLK 4, CLK 2, CLK 1, CLK 512 andCLK 256 are each derived from CLK 16 by a programmable division, in thiscase by an even power of 2. These clocks operate at 8 MhZ, 4 MhZ, 1 MhZ,512 KhZ and 256 KhZ, respectively. In addition, an S clock signal isderived and a clock sync signal CLKSYNC occurs every 8 milaseconds whichresets the clock dividers and ensures that all clocks operate in lock. Aplurality of ADC clocks are shown. These clocks may be, for example,clocks associated with the ADC interface 110 shown in FIG. 1. They areutilized for controlling whatever operations might be desirable withinthat circuit. In this case, a plurality of different clocks are shown.However, what is important is that each of these clocks utilized withoff chip devices are generated on the falling edge of CLK 16. Thus, theactivities which occur on the chip shown in FIG. 7 will occur atdifferent instances from the activities occurring on external devices.This provides considerable advantage when dealing with noise and otherdesign issues. The synchronization of clocks on a chip, in this case forexample on the RSU chip is particularly advantageous because it easesthe interfacing of on chip components because of the known timerelationships.

FIG. 29 is a block diagram showing how clock reprogramability isimplemented in accordance with the invention. This process is describedin conjunction with FIG. 28 in which a 16 megabit per second chip clockis provided to a programmable divider 2900 which divides the clock downto a local chip clock frequency 2910. A register 2920 is connected tothe TMI bus 705 so that the value in the register 2920 can be programmedfrom the TMI bus. However, the revised value in the register 2920 cannothe applied to the programmable divider 2900 until the occurrence of async pulse 2930.

By switching the programming of a clock during the sync pulse, the clockcan be reprogrammed during operation without cause causing glitches inthe data. Further, data interfacing among devices on the chip is easierwhen all clocks on the chip are synchronized.

A problem exists when implementing mathematics in the DSP. The problemis that many adder circuits do not correctly determine a carry bit. Inaccordance with the invention, a carry detection circuit has beendeveloped which can detect correctly the carry bit ofX*Y+Accumulator+round. X*Y+Accumulator has been called MACtraditionally. Previous work has been addressed to X*Y+Accumulator.However, with rounding, the circuit is not obviously correct and is, infact, many times incorrect because the intimediate values are scrambled.The carry detection circuit described here overcomes this problem.

The following 5 steps are undertaken in order to determine the carry bitcorrectly.

1. Determine if product is negative.

2. Determine if accumulator is negative.

3. Determine if the round-bit propagates all the way to the mostsignificant bit, MSB.

4. Determine if result (X*Y+Accumulator+round) is negative.

5. Determine a correct carry bit (based on previous 4 results).

The actual circuit implementation of the previous steps are described asfollows.

1. negative product bit: (proof 1)

(multinA[MSB] {circumflex over ( )} multinB[MSB]) && |multinB &&|multinA

multinA: an N-bit 2's complement number

multinB: an N-bit 2's complement number

MSB: “Most Significant Bit” i.e. bit N−1

Note: one counts bit 0, bit1, . . . bit N−1.

Thus, the number of bits is equal to N, but the most significant one isbit N−1.

{circumflex over ( )}: logical XOR operation

&& : logical AND operation

|: bitwise logical OR operation

e.g. |multinB means multinB[N−1] OR multinB[N−2] OR . . .

. . . OR multinB[0]

2. negative accumulator bit:

acc[MSB]

acc: 2's complement number of Accumulator

Note that acc has>2N bits to store results of previous multiplications.

e.g. 1010*0101=11100010 thus, 4-bit number*4-bit number becomes 8-bitnumber.

It is a property of 2's complement number that the MSB is the sign bit.

3. round-bit propagates to MSB bit: (proof 3)

Let i be the bit that round is added to accumulator output

rndprop (round-propagate) bit:

round && (result[MSB:I] all zero)

result: X*Y+Accumulator+round

round: user can choose to round or not. 1 means yes,

0 means no

i: usually is bit N−1

e.g. 1010*0101+11000011+00001000

ace round

N=4, thus, 4 bit operands, acc has 8 bits, and round is added at bit 3(i.e. N−1).

4. negative result bit:

result[MSB]

result : X*Y+Accumulator+round

5. (x is don't care) (proof 5)

casex ({sign_Product, sign_Acc, sign_Result, rndprop})

4′b0000: cout <=0;

4′b0001: cout <=1;

4′b001x: cout <=0;

4′b010x: cout <=1;

4′b0110: cout <=0;

4′b0111: cout <=1;

4′b100x: cout <=1;

4′b1010: cout <=0;

4′b1011: cout <=1;

4′b110x: cout <=1;

4′b111x: cout <=1;

endcase

sign_Product: negative product bit from 1.

sign_Acc: negative accumulator bit from 2.

sign_Result: negative result bit from 4.

Rndproop : round-bit propagate to MSB bit from 3.

Note: There should be 2 carry bits (proof 5).

However, as implemented they are logically ORed together, just to makeit fit the traditional circuit.

The following are 2 examples which illustrates 3 and 5. Finally, theproof for 1, 3, and 5 are provided.

The area of this carry detection circuit, as in proof 5, is:

1 nr2 (2 p 2 n)

3 inv (1 p 1 n)

ao21 (4 p 4 n)

oai221l (5 p 5 n)

Total: 6 logic gates 14 p 14 n

nr2: logical 2 input NOR gate, i.e.˜(J∥K)

inv: logical inverter i.e. ˜J

ao21: logical 2 input AND-OR, i.e. (J∥K)∥

L

oai221: logical OR-AND-INV i.e. ˜((J∥K) && (L∥M))

The area of 1, 3, can be shared with different overflow, and zerodetection circuit, which is usually in place with the carry out circuit.

EXAMPLES

Here is a brief examples of how 3, and 5 works.

Example for Proof 3

The following are all binary numbers:

(0 is zero, 1 is one, X is don't care)

One can deduce that the carryout from the leftmost bit is 1.

Explanation: if one adds 1 at bit k and gets 0 at the output, one knowsthat there is a carryout to the next bit (k+1) location. Again, if oneadds that carry to bit k+1 and get a 0 at the output, one knows thatthere is a carryout to the next bit (k+2) location. Similar, one cancontinue on and on, thus deduct that there is a carryout from theleftmost bit.

Example for Proof 5

All numbers are 2's complement binary numbers

Suppose one adds two numbers and rounds.

The circuit in previous work does not address the previous situationscorrectly.

Although the present invention has been described and illustrated indetail, it is clearly understood that the same is by way of illustrationand example only and is not to be taken by way of limitation, the spiritand scope of the present invention being limited only by the terms ofthe appended claims and their equivalents.

What is claimed is:
 1. A linear phase FIR sinc filter, comprising: a. abus; b. one or more input buffers, one for each incoming channelconnected to said bus; c. one or more pages of memory, one for eachinput buffer, connected to said bus; d. a shifter, connected to saidbus, selectively shifting the bit positions of digital words receivedfrom said bus; and e. an arithmetic unit receiving at one input theoutput of said shifter and at another input an output of an accumulatorand selectively providing an output either to said bus or to a filteroutput or to an input of said accumulator.
 2. The linear phase FIR sinefilter of claim 1, further comprising a sequencer for controllingswitching of said sinc filter.
 3. The linear phase FIR sinc filter ofclaim 2 in which said sequencer is partitioned into main and subroutinecomponents.
 4. The linear phase FIR sinc filter of claim 3 in whichinvocation of a component of one of said main components invokesexecution of one or more subroutine components.
 5. The linear phase FIRsinc filter of claim 3 in which said sequencer is implemented inmicrocode.
 6. The linear phase FIR sinc filter of claim 3 in which saidsequencer is implemented in logic.
 7. A method of implementing a sincfilter, comprising the steps of: a. receiving a plurality of digitalwords; b. selectively shifting one or more of said words; c. selectivelyadding previously stored results to said shifted one or more of saidwords to produce a summed output; and d. selectively storing said summedoutput or providing said summed output to a filter output or to anaccumulator input.
 8. A program product, comprising: a. a memory medium;and b. control instructions stored on said memory medium, said controlinstructions causing a plurality of digital words to be received,selective shifting one or more of said words, selective adding ofpreviously stored results to one or more shifted words to produce asummed output; and selectively storing or outputting of said summedoutput.